Hybrid power cell

ABSTRACT

The present invention relates to a bi-directional power converter (100) for converting an input voltage into an output voltage. The bi-directional power converter (100) may comprise a primary switching unit having a primary switching frequency and a secondary switching unit having a secondary switching frequency: the primary switching frequency being lower than the secondary switching frequency.

TECHNICAL FIELD OF THE INVENTION

The present invention relates to a power converter for converting aninput power into an output power, a boost process and/or a buck processfor converting an input power into an output power. More particularly,the present invention is dedicated, but not exclusively, to electricaltraction systems able to drive electrical motors and to recover kineticenergy towards the embedded Energy Storage Systems.

STATE OF THE ART

As the technologies of batteries and supercapacitors were improved overthe last decades, electric vehicles have become an attractive solutionwith an economic reality. Moreover, due to the commitment of severalcountries around the world to respect the COP21 act, traditional fullgasoline vehicle may be abandoned replacing by electrical solution. Oneof the first application will be the public transport like bus andtramway systems for example. In these applications, high power isrequired to supply the traction units.

This involves a parallel connection of supercapacitors and batteriesthat is more generally called Energy Storage Elements, also know as ESE.This electrical configuration can be achieved thanks to a high powerconverter (100) that manages energy between traction units and severalESEs. This electrical solution is distinguished by its ability torecover the kinetic energy, so that allows improving significantly theglobal system efficiency. These criteria may involve using abidirectional high power converter with a high efficiency.

These bidirectional high power converters work at high voltage and therisk of faults occurring on the power semiconductor, at this voltagelevel, is linked to the cosmic rays effect depends on altitude and DCbias voltage. For example, the MTBF, an acronym for “Mean Time BetweenFailure”, for one high voltage component may be around 15 years for somemarket whereas it should be higher than 30 years for railwayapplications. Thus, the FIT value, short for “Failure In Time”, due tothe cosmic rays effect from power semiconductor should be well analyzedversus the final applications. Sometime it is mandatory to use highervoltage breakdown voltage to respect the MTBF expectation. It is obviousthat using 1.7 kV Insulated Gate Bipolar Transistors, IGBT for short,instead 1.2 kV IGBT have a huge consequence on the switching frequencyfor the same power losses. This involves a consequence on the size andthe weight of the systems.

An alternative may be the use of Silicon carbide power modules, SiCpower modules for short, which are often considered to improveefficiency and size of the power converter. However, these componentsare new on the market and there are no feedbacks on their ability towork under a large temperature range over a long time period especiallyin the applications in which there are thermal cycling constraints.Moreover, their cost remains higher than the standard Silicon powermodules. To address the market of the public electrical vehicles forexample, needing high current repetitive quick charge, under acompetitive market, full SiC power modules cannot be yet considered.

Indeed, due to the random high power cycles, the thermal cycling of thesemiconductor should be considered during the power converter design.Those thermal stresses are due to the quick charges from the kineticenergy recovery, during braking phase, due to the high sourcing inrushcurrent required by the traction units, and/or due to the low powerconsumption from the Heating Ventilation Air Conditioning for instance.

To reduce joule losses in the electrical wires connections, serialcombination of the ESE is used, involving a high voltage, around 0.9 kV,across the power converter. This voltage leads a specific attentionabout the life time of the semiconductor due to the combination of thecosmic ray versus the junction operating voltage; the immunity of theSiC and Si components should be study to not affect the reliability ofthe system which become crucial for land transport as bus, tramways,trains, air transport such as aircraft, and sea transport like boat.

As power converter is embedded in rolling stock, size and weight shouldbe considered to find the best electrical topology. As a consequence, ahigh switching frequency topology will be considered.

On top of that, in public transport system particularly, the quickrecharges of the ESEs occur during brief stops when the passengers go upand down which can generate a high audible noise due to themagnetrostriction effect of the magnetic elements and may annoy thepassengers.

In several soft switching power converters, it is often impossible towork without any loads because the snubber capacitors are fastdischarged through the power semiconductors. This loss reduces thelifetime of the system.

SUMMARY OF THE INVENTION

In order to achieve this objective, the present invention may provide,according to a aspect of the invention, a boost or buck power converter(100) for converting a input primary power having a input primaryvoltage (201) and a input primary current (202) into a output secondarypower having a output secondary voltage (203) and output secondarycurrent (204) or a input secondary power having said input secondaryvoltage (291) and a input secondary current (292) into a output primarypower having a output primary voltage (293) and second primary current(294) respectively; said power converter (100) including at least one:

-   -   primary switching arrangement (130) comprising at least one        primary switching unit (131, 135) having a primary dynamical        switching loss;    -   secondary switching arrangement (170) comprising at least one        secondary switching unit (171, 175) having a secondary dynamical        switching loss; and,    -   resonant unit (150) connecting said at least one primary        switching unit (131, 135) and said at least one secondary        switching unit (171, 175);

said primary dynamical switching loss being higher than said secondarydynamical switching loss.

Thus, this arrangement improves the efficiency, the lifetime, thereliability, the thermal resistance of the primary arrangement versusthe secondary arrangement and the audible noise while reducing the size.Further, this arrangement improves the switching frequency of theprimary switching arrangement (130) and increases the chopping frequencyof the boost or buck power converter (100). Moreover this arrangementreduces the voltage slope applied the external magnetic elements (111)compared to the traditional hard switching improving their lifetime andreducing the electromagnetic interference.

By primary dynamical switching loss, we mean a function of the turn-onenergy, the turn-off energy and/or the reverse recovery energy of theprimary switching unit characterized in a hard-switching mode. Inparticular, the primary dynamical switching loss may be defined as a sumof the turn-on energy, the turn-off energy and/or the reverse recoveryenergy of the primary switching unit.

By secondary dynamical switching loss, we mean a function of the turn-onenergy, the turn-off energy and/or the reverse recovery energy of thesecondary switching unit characterized in a hard-switching mode. Inparticular, the secondary dynamical switching loss may be defined as asum of the turn-on energy, the turn-off energy and/or the reverserecovery energy of the secondary switching unit.

The power converter (100) may comprise one or a plurality of thefollowing technical features which may be considered alone or incombination.

According to an embodiment, said resonant unit (150) may comprise atleast one inductor (153) and one capacitor (151, 152).

Thus, this arrangement allows adjusting the resonance of the resonantunit (150).

According to an embodiment, said primary switching arrangement (130) maycomprise at least a first primary switching unit (131) and a secondprimary switching unit (135) and/or wherein said secondary switchingarrangement (170) may comprise at least a first secondary switching unit(171) and a second secondary switching unit (175).

According to an embodiment, a dimension of primary switching arrangement(130) may be larger than a dimension of the secondary switchingarrangement (170). According to an embodiment, the size of primaryswitching arrangement (130) may be bigger than the size of the secondaryswitching arrangement (170).

By dimension, we mean a component dimension.

Thus, this arrangement the primary switching arrangement (130) maydissipated more heat than the secondary switching arrangement (170) andthe reliability of the primary switching arrangement (130) may be betterthan the secondary switching.

According to an embodiment, the primary switching arrangement (130) maycomprise a primary dynamical switching loss and said secondary switchingarrangement (170) may comprise a secondary dynamical switching loss;said primary dynamical switching loss may be higher than said secondarydynamical switching loss.

Thus, this arrangement reduces the audible noise.

According to an embodiment, said at least one primary switching unit(131, 135) may be a Gate Turn-OFF thyristor, Insulated-Gate BipolarTransistor, Field Effect Transistor and/or Metal-Oxide-SemiconductorField-Effect Transistor and/or said at least one secondary switchingunit (171, 175) may be a Field Effect Transistor and/or aMetal-Oxide-Semiconductor Field-Effect Transistor.

Thus, this arrangement linked of the thermal resistance losses andimproves the lifetime, the thermal cycle, the reliability and the sizeof the power converter (100).

According to an embodiment, the power converter (100) may comprise atleast one:

-   -   primary terminal (110): said primary terminal (110) may be        configured to connect a primary electrical component 901 to said        power converter (100);    -   secondary terminal (190): said secondary terminal (190) may be        configured to connect a secondary electrical component 902 to        said power converter (100);    -   common terminal (145): said common terminal (145) may be        connected to said primary electrical component 901 to said        secondary electrical component 902;

said primary switching arrangement (130) may be connected to saidprimary terminal (110), to said secondary terminal (190), and to saidcommon terminal (145), and said secondary switching arrangement (170)may be to connected to said secondary terminal (190) and to said commonterminal (145).

Thus, this arrangement allows the power converter (100) to be connecteda load and an energy source.

According to an embodiment, the primary switching arrangement (130) maycomprise at least a first primary switching unit (131) and a secondprimary switching unit (135) and/or wherein the secondary switchingarrangement (170) may comprise at least a first secondary switching unit(171) and a second secondary switching unit (172).

According to an embodiment, the primary switching arrangement (130) maycomprise a primary dynamical switching loss and said secondary switchingarrangement (170) may comprise a secondary dynamical switching loss;said primary dynamical switching loss may be higher than said secondarydynamical switching loss.

Thus, this arrangement reduces the audible noise.

According to an embodiment, the primary switching frequency may be maycomprise between 2 kHz and 60 kHz, in particular between 2.9 kHz and 53kHz, and preferably between 3.4 kHz and 46 kHz and said secondaryswitching frequency may be may comprise between 5 kHz and 96 kHz, inparticular between 9 kHz and 83 kHz, and preferably between 14 kHz and71 kHz.

Another aspect of the present invention may provide a boost or buckprocess (501, 502) for converting a input primary power having a inputprimary voltage (201) and a input primary current (202) into a outputsecondary power having a output secondary voltage (203) and outputsecondary current (204) or a input secondary power having said inputsecondary voltage (291) and a input secondary current (292) into aoutput primary power having a output primary voltage (293) and secondprimary current (294) respectively; said boost or buck process (501,502) comprising at least a step of:

-   -   providing a power converter (100) according to claims 1 to 6;    -   switching OFF of at least one secondary switching unit (171,        175) when a voltage across said at least one primary switching        unit (131, 135) reaches a predetermined ratio of said output        secondary voltage (203) or said output primary voltage (293)        respectively.

According to an embodiment, said predetermined ratio of said outputsecondary voltage (203) or said output primary voltage (293)respectively may be may comprise between 20% and 80%, in particularbetween 40% and 60% and preferably between 45% and 55%.

According to an embodiment, said voltage across said at least oneprimary switching unit (131, 135) may be a voltage across said at leastone capacitor (151, 152).

According to an embodiment, said at least one secondary switching unit(171, 175) may comprise intrinsic characteristic controlling the time ofsaid switching OFF step.

According to an embodiment, said intrinsic characteristic may be a gateresistor of said at least one secondary switching unit (171, 175).

According to an embodiment, said voltage across said at least oneprimary switching unit (131, 135) reaches a voltage between saidsecondary terminal (190) and said primary terminal (110).

According to an embodiment, said voltage across said at least oneprimary switching unit (131, 135) reaches a voltage between said primaryterminal (110) and said common terminal (145).

According to an embodiment, a current of said resonant unit (150)reaches a maximum resonant current.

According to an embodiment, the boost or buck process (501, 502) maycomprise a switch ON step of said at least one primary switching unit(131, 135) when said voltage across said at least one primary switchingunit (131, 135) reaches a minimum voltage.

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   a direct first primary current 233 (IDH) flows through the first        primary switching unit from the primary terminal to the        secondary terminal and reaches a minimum first primary current;    -   a direct second resonant voltage (VCL) measured between the        primary terminal and the common terminal drops;    -   a direct resonant current 251 (ILR) flows from the primary        terminal to the common terminal through the resonant unit and        reaches a maximum direct resonant current 251; and,    -   a reverse second primary current 236 (IQL) flows through the        second primary switching unit from the primary terminal to the        common terminal and may be different than a reference reverse        second primary current 236.

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct first primary current 233 (IDH) reaches a reference        first primary current;    -   said direct second resonant voltage (VCL) reaches a reference        direct second resonant voltage;    -   said direct resonant current 251 (ILR) drops; and,    -   said reverse second primary current 236 (IQL) reaches a minimum        reverse second primary current 236.

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct first primary current 233 (IDH) stays at said        reference first primary current;    -   said direct second resonant voltage (VCL) stays at said        reference direct second resonant voltage;    -   said direct resonant current 251 (ILR) reaches a reference        direct resonant current 251; and,    -   said reverse second primary current 236 (IQL) reaches a maximum        reverse second primary current 236.

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct first primary current 233 (IDH) stays at said        reference first primary current;    -   said direct second resonant voltage (VCL) increases;    -   said direct resonant current 251 (ILR) stays at said reference        direct resonant current 251; and,    -   said reverse second primary current 236 (IQL) drops.

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct first primary current 233 (IDH) reaches a maximum        first primary current;    -   said direct second resonant voltage (VCL) reaches a maximum        direct second resonant voltage;    -   said direct resonant current 251 (ILR) flows from the primary        terminal to the secondary terminal through the resonant unit;        and,    -   said reverse second primary current 236 (IQL) reaches said        reference reverse second primary current 236;

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct first primary current 233 (IDH) stays at said        maximum first primary current;    -   said direct second resonant voltage (VCL) stays at said maximum        direct second resonant voltage;    -   said direct resonant current 251 (ILR) flows from the primary        terminal to the secondary terminal through the resonant unit;        and,    -   said reverse second primary current 236 (IQL) stays at said        reference reverse second primary current 236;

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct first primary current 233 (IDH) stays at said        maximum first primary current;    -   said direct second resonant voltage (VCL) stays at said maximum        direct second resonant voltage;    -   said direct resonant current 251 (ILR) flows from the primary        terminal to the secondary terminal through the resonant unit;        and,    -   said reverse second primary current 236 (IQL) stays at said        reference reverse second primary current 236;

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct first primary current 233 (IDH) stays at said        maximum first primary current;    -   said direct second resonant voltage (VCL) stays at said maximum        direct second resonant voltage;    -   said direct resonant current 251 (ILR) flows from the primary        terminal to the secondary terminal through the resonant unit;        and,    -   said reverse second primary current 236 (IQL) stays at said        reference reverse second primary current 236.

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   a direct second primary current 237 (IDL) flows through the        second primary switching unit from the common terminal to the        primary terminal and reaches a minimum second primary current;    -   a direct first resonant voltage (VCH) measured between the        primary terminal and the secondary terminal drops;    -   a second resonant current (ILR) flows from the secondary        terminal to the primary terminal through the resonant unit and        reaches a maximum direct resonant current 251; and,    -   a reverse first primary current (IQH) flows through the first        primary switching unit from the secondary terminal to the        primary terminal and may be different than a reference reverse        first primary current;

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct second primary current 237 (IDL) reaches a reference        first secondary current;    -   said direct first resonant voltage (VCH) reaches a reference        direct first resonant voltage;    -   said second resonant current (ILR) drops; and,    -   said reverse first primary current (IQH) reaches a minimum        reverse first primary current;

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct second primary current 237 (IDL) stays at said        reference first secondary current;    -   said direct first resonant voltage (VCH) stays at said reference        direct first resonant voltage;    -   said second resonant current (ILR) reaches a reference second        resonant current; and,    -   said reverse first primary current (IQH) reaches a maximum        reverse first primary current;

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct second primary current 237 (IDL) stays at said        reference first secondary current;    -   said direct first resonant voltage (VCH) increases;    -   said second resonant current (ILR) reaches a reference second        resonant current; and,    -   said reverse first primary current (IQH) drops;

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct second primary current 237 (IDL) reaches a maximum        direct first secondary current;    -   said direct first resonant voltage (VCH) reaches a maximum        direct first resonant voltage;    -   said second resonant current (ILR) flows from the secondary        terminal to the primary terminal through the resonant unit; and,    -   said reverse first primary current (IQH) reaches said reference        reverse first primary current;

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct second primary current 237 (IDL) stays at said        maximum direct first secondary current;    -   said direct first resonant voltage (VCH)) stays at said maximum        direct first resonant voltage;    -   said second resonant current (ILR) flows from the secondary        terminal to the primary terminal through the resonant unit; and,    -   said reverse first primary current (IQH) stays at said reference        reverse first primary current;

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct second primary current 237 (IDL) stays at said        maximum direct first secondary current;    -   said direct first resonant voltage (VCH)) stays at said maximum        direct first resonant voltage;    -   said second resonant current (ILR) flows from the secondary        terminal to the primary terminal through the resonant unit; and,    -   said reverse first primary current (IQH) stays at said reference        reverse first primary current;

According to an embodiment, the boost or buck process (501, 502) maycomprise a step:

-   -   said direct second primary current 237 (IDL) stays at said        maximum direct first secondary current;    -   said direct first resonant voltage (VCH)) stays at said maximum        direct first resonant voltage;    -   said second resonant current (ILR) flows from the secondary        terminal to the primary terminal through the resonant unit; and,    -   said reverse first primary current (IQH) stays at said reference        reverse first primary current.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing and other purposes, features, aspects, and advantages ofthe invention will become apparent from the following detaileddescription of embodiments, given by way of illustration and notlimitation with reference to the accompanying drawings, in which:

FIG. 1 represents buck power converter according to an embodiment of thepresent invention;

FIG. 2 shows boost power converter according to the same embodiment ofthe present invention;

FIGS. 3 to 17 illustrate the sequences of the boost process according tothe same or another embodiment of the present invention;

FIGS. 18 to 32 present the sequences of the buck process according toanother embodiment of the present invention;

FIG. 33 represents a control sequences; and,

FIG. 34 illustrates

DESCRIPTION OF THE INVENTION

The present invention may provide a power converter 100 with an improvedlifetime, and reliability while reducing the size and the audible noise.The power converter 100 of the present invention, as depicted on FIGS. 1and 2, may be deemed to improve the lifetime of Energy Storage Systems,ESS for short, also known as Energy Storage Elements, ESE for short, byminimizing current flowing through them. The power converter 100described in this present application may be dedicated to energy storagesystems having a high nominal voltage, which may mean comprised between750 VDC and 950 VDC. In particular, the present invention may provide abi-directional power converter 100 since the power converter 100 mayfunction as a boost converter 100 or a buck converter 100.

This power converter 100 may include at least one primary switchingarrangement 130, secondary switching arrangement 170 and a resonant unit150 connecting the primary switching arrangement 130 and the secondaryswitching arrangement 170.

The primary switching arrangement 130 may comprise at least one primaryswitching unit 131, 135 having a primary switching frequency. In thepresent embodiment, the primary switching arrangement 130 may compriseat least a first primary switching unit 131 including a first primarytransistor 132 and a first primary diode 133 and a second primaryswitching unit 135 including a second primary transistor 136 and asecond primary diode 137. The first and second primary transistors 131,136 may be a Gate Turn-OFF thyristor, Field Effect Transistor,Metal-Oxide-Semiconductor Field-Effect Transistor and/or anInsulated-Gate Bipolar Transistor, IGBT for short.

The secondary switching arrangement 170 may comprise at least onesecondary switching unit 171, 175 having a secondary switchingfrequency. In the present embodiment, the secondary switchingarrangement 170 may comprise at least a first secondary switching unit171 including a first secondary transistor 172 and a first secondarydiode 173 and a second secondary switching unit 175 including a secondsecondary transistor 176 and a second secondary diode 177. The first andsecond secondary transistors 172, 176 may be a Field Effect Transistorand/or a Metal-Oxide-Semiconductor Field-Effect Transistor, MOSFET forshort. MOSFET and in particular the Silicon CarbideMetal-Oxide-Semiconductor Field-Effect Transistor, SiC MOSFET for short,may have been selected to avoid the recovery charges and the tailcurrents met with Silicon transistor like Silicon IGBT. This selectionof SiC MOSFET may optimize the benefits of a soft switching process.

Another way to distinguish the primary switching arrangement 130 andsecondary switching arrangement 170 may be the size or dimension of thecomponent, since the dimension of the primary switching arrangement 130may be larger than the size of the secondary switching arrangement 170.A larger dimension of the primary switching arrangement 130 improves theheat dissipation and reliability of the primary switching arrangement130 since reliability may be linked with the thermal capacity of theprimary switching arrangement 130.

The primary switching unit 130 may comprise a primary dynamicalswitching loss and said secondary switching unit 170 may comprise asecondary dynamical switching loss.

The primary dynamical switching loss may be defined as a function of theturn-on energy, the turn-off energy, and/or the reverse recovery energyof the primary switching unit. In particular, the primary dynamicalswitching loss may be defined as a sum of the turn-on energy, theturn-off energy, and/or the reverse recovery energy of the primaryswitching unit.

The turn-on energy, the turn-off energy, and/or the reverse recoveryenergy may be characterized in hard-switching mode.

The primary switching unit 130 differs from said secondary switchingunit 170 in that a small direct drop voltage when high current flowsthrough it, e.g. 1.5V for 110 A, but it has high dynamical switchingpower losses during the turn ON E_(ON1) and during the turn OFFE_(OFF1), and the turn OFF E_(RR1) of the diode function.

The primary switching unit 130 differs also from said secondaryswitching unit 170 in that a surface area of semiconductor material ofthe primary switching unit 130 is greater said secondary switching unit170, and the cost in square millimeter of the primary switching unit130, ca. 0.06 €/mm², is lower than the cost in square millimeter of thesecondary switching unit 170, ca. 1.8 €/mm².

The turn ON E_(ON1) is characterized by the product of electricalcurrent/voltage quantities at the turn ON, for a hard switching, of theprimary switching unit 130.

The turn OFF E_(OFF1) is characterized by the product of thecurrent/voltage electrical quantities at the turn OFF, for a hardswitching, of the primary switching unit 130.

The turn OFF E_(RR1) of the diode function is characterized by theproduct of the current/voltage electrical quantities at the turn OFF,for a hard switching, of the primary switching unit 130.

The turn-on energy may be comprised 15 mJ and 153 mJ. The turn-offenergy may be comprised 17 mJ and at least 193 mJ. The recovery reverseenergy may be comprised 16 mJ and 163 mJ.

For example, for high power applications, the primary switching unit130, slow during the dynamical switching, will be a 1.2 kV IGBT or a 1.7KV IGBT. In this application, the dynamical switching energies the turnON E_(ON1) are equal to 22 mJ for the 1.2 kV IGBT and 53 mJ for the 1.7kV IGBT. The turn OFF E_(OFF1) dynamical switching energies are equal to27 mJ for the 1.2 kV IGBT and 58 mJ for the 1.7 kV IGBT. Finally, thedynamical switching energies of the turn OFF E_(RR1) of the diodefunction are equal to 28 mJ for the 1.2 kV IGBT and 55 mJ for the 1.7 kVIGBT.

The surface area of the primary switching unit 130 is typically close to450 mm² for a direct current of 110 A and its cost is equal to 0.06€/mm².

The secondary switching unit 170 is distinguished from the primaryswitching unit 130 by a high direct drop voltage when high current flowsthrough it, e.g. 5V for 110 A, but it has, intrinsically, low powerdynamical switching losses during the turn ON E_(ON2) and during theturn OFF E_(OFF2) and the turn OFF E_(RR2) of the diode function of thesecondary switching unit 170.

For example, for high power applications, the dynamical fast switchingbehavior of the secondary switching unit 170 is either a SiC 1.2 kVMOSFET or a 1.7 KV SiC MOSFET. In this application the dynamicalswitching energies of the turn ON E_(ON2) are equal to 3.3 mJ for theSiC 1.2 kV MOSFET and 6 mJ for the 1.7 kV SiC MOSFET. The dynamicalswitching energies of the turn OFF E_(OFF2) are equal to 1.8 mJ for theSiC 1.2 kV MOSFET and 3 mJ for the 1.7 kV SiC MOSFET. Finally, thedynamical switching energies of the turn OFF E_(RR2) of the diodefunction of the secondary switching unit 170 are equal to 180 μl for theMOSFET 1.2 kV and 240μJ for the MOSFET 1.7 kV.

The surface area of the semiconductor of the secondary switching unit170 is close to 36 mm² for a current switched close to 110 A and itscost is typically equal to 1.8 €/mm².

The primary switching frequency of the primary switching unit may be afunction of said primary switching loss. This primary dynamicalswitching frequency may be may comprise between 2 kHz and 60 kHz, inparticular between 2.9 kHz and 53 kHz, and preferably between 3.4 kHzand 46 kHz.

On other side, the secondary dynamical switching loss may be defined asa function of the turn-on energy, the turn-off energy, and/or thereverse recovery energy of the secondary switching unit. In particular,the secondary dynamical switching loss may be defined as a sum of theturn-on energy, the turn-off energy, and/or the reverse recovery energyof the secondary switching unit. The turn-on energy, the turn-offenergy, and/or the reverse recovery energy may be characterized inhard-switching mode.

The turn-on energy may be comprised between 0 and 100 mJ. The turn-offenergy may be comprised between 100 μJ and at least 1 mJ. The recoveryreverse energy may be comprised between zero and 1 mJ.

The secondary switching frequency of the secondary switching unit may bea function of said secondary switching loss. This secondary dynamicalswitching frequency may be comprised between 5 kHz and 96 kHz, inparticular between 9 kHz and 83 kHz, and preferably between 14 kHz and71 kHz.

As it may be understood, the primary switching frequency may be lowerthan the secondary switching frequency.

When the primary switching arrangement 130 may comprise more than oneswitching unit, the primary dynamical switching loss of the primaryswitching arrangement 130 may be linked with the first primary switchingfrequency of the first primary switching unit 131, in particular of thefirst primary transistor 132, and the second primary switching frequencyof the second primary switching unit 135, in particular of the secondprimary transistor 136.

When the secondary switching arrangement 170 may comprise more than oneswitching unit, the secondary dynamical switching loss of the secondaryswitching arrangement 170 may be linked with the first secondaryswitching frequency of the first secondary switching unit 171, inparticular of the first secondary transistor 172, and the secondsecondary switching frequency of the second secondary switching unit175, in particular of the second secondary transistor 176.

As pictured in FIG. 1, the power converter 100 may comprise a primaryterminal 110, a second terminal 190 and a common terminal 145. Theprimary terminal 110 may comprise a primary inductor 111 and/or aprimary capacitor 112, and the secondary terminal 190 may comprise asecondary capacitor 191.

The primary terminal 110 may be configured to connect a primaryelectrical component 901, having input primary voltage of V₂₀₁ or outputprimary voltage of V₂₉₃, which may be a load and/or an energy source, tothe power converter 100, the secondary terminal 190 may be configured toconnect a secondary electrical component 902, having output secondaryvoltage of V₂₀₃ or input secondary voltage of V₂₉₁, which may be a loadand/or an energy source, to the power converter 100 and the commonterminal 145 may be connected to the primary electrical component 901and to the secondary electrical component 902.

On FIG. 1, the primary switching arrangement 130 may be connected to theprimary terminal 110, to the secondary terminal 190, and to the commonterminal 145. As previously mentioned, the primary switching arrangement130 may comprise the first primary switching unit 131 connecting theprimary terminal 110 and the secondary terminal 190, and the secondprimary switching unit 135 connecting the primary terminal 110 and thecommon terminal 145. In particular, the first primary switching unit 131may connect the primary terminal 110 and the secondary terminal 190 viathe first primary transistor 132 and/or the first primary diode 133, andthe second primary switching unit 135 connecting the primary terminal110 and the common terminal 145 via the second primary transistor 136and/or the second primary diode 137.

The secondary switching arrangement 170 may be connected to thesecondary terminal 190 and to the common terminal 145. The secondaryswitching arrangement 170 may comprise the first secondary switchingunit 171 connecting the secondary terminal 190 and the resonant unit150, and the second secondary switching unit 175 connecting the resonantunit 150 and the common terminal 145. In particular, the first secondaryswitching unit 171 may connect the secondary terminal 190 and theresonant unit 150 via the first secondary transistor 172 and/or thefirst secondary diode 173, and the second secondary switching unit 175connecting the resonant unit 150 and the common terminal 145 via thesecond secondary transistor 176 and/or the second secondary diode 177.

The resonant unit 150, mentioned above, may be connected to the primaryterminal 110, to the secondary terminal 190, and to the common terminal145. The resonant unit 150 may comprise a first capacitor 151 connectingthe primary terminal 110 and the secondary terminal 190, and a secondcapacitor 152 connecting the primary terminal 110 and the commonterminal 145. An inductor 153 of the resonant unit 150 may connect theprimary terminal 110 and the first and second secondary transistors 172,176 and/or the first and second secondary diodes 173, 177.

In some embodiment, the resonant unit 150 may comprise at least oneinductor 153 connecting the primary terminal 110 and the first andsecond secondary transistors 172, 176 and/or the first and secondsecondary diodes 173, 177 and one capacitor 151, 152 connecting theprimary terminal 110 and the secondary terminal 190 or the primaryterminal 110 and the common terminal 145.

The boost or buck process may have been calculated to reduce the numberof first and second secondary switching units 171, 175 chips and todecrease the thermal cycling stress.

Inductor 153 has been chosen to control the time rate of change ofintensity (di/dt) at each commutation around 50 A/μs 20 μH, reducingdrastically the power losses due to recovery charge of the firstsecondary diode 173 and second secondary diode 177 embedded inside thesecondary switching arrangement 170.

The boost or buck process may have been calculated in order to turns ONat zero voltage and it turns OFF under zero voltage the primaryswitching arrangement 130.

First and second Capacitors 151, 152 may reduce the tail power lossesand fix time rate of change of potential across the primary andsecondary electrical components 901, 902 around 1 kV/μs. The time rateof change of potential of the primary switching arrangement 130 may be20 times lower than the secondary switching arrangement 170 working inhard switching, reducing the electromagnetic interference effects, EMIfor short.

Description of a Boost Process

The power converter, according the present invention, may be abidirectional power converter 100, also called bidirectional power cell.It may work like Boost or Buck. The power converter 100 may be used forthe power inverters or for the battery charger dedicated to the tractionsystem for instance.

The description below shows the sequence when the power converter 100works like a boost power converter 100 and implementing a boost process,which may mean that an energy source 901 may be connected to the primaryterminal 110 and the common terminal 145 and a load 902 may be connectedto the secondary terminal 190 and the common terminal 145. For readingconvenience and better understanding, the switching and resonant units130, 150, 190, which may be OFF, may be dotted and others, which may beON, may be draw with solid line.

In the present invention, by “OFF” and “no current” we mean that almostno current may flow, a current lower or equal than 1 mA or the flowingcurrent may be not intended. On other side, by “ON” we mean that acurrent may flow, a current higher than 1 mA or the flowing current maybe intended.

Also, the terms direct and reverse may be just used for readingconvenience and may indicate upward and downward respectively and rightand left respectively in the direction of the figure.

Further, by the terms “load”, we mean all electronic/electrical deviceswhich may transform or consume electrical energy like a motor, abattery, a supercapacitor or a resistor, but not limited thereto, and by“source” we mean all energy suppliers which may supply electrical energylike a motor, a battery, or a supercapacitor for instance, but notlimited thereto.

During an initial step, an input primary current 202 flows through theprimary inductor 111, having an inductance L₁₁₁, and the primaryswitching arrangement 130 from the energy source to the secondaryterminal 190. In particular, the input primary current 202 flows to thesecondary terminal 190 through the primary inductor 111 and at least oneof a primary switching unit comprised in the primary switchingarrangement 130 and preferably in the first primary switching unit 131.No current flows in the others units, which mean no current flowsthrough a second primary switching unit 135 may comprise the primaryswitching unit, the resonant unit 150 and the secondary switchingarrangement 170.

As depicted in FIG. 3, a direct second primary current 237 flows throughthe second primary diode 137 of the second primary switching unit 135from the common terminal 145 to the primary terminal 110.

After this initial step, a first boost step occurs wherein the secondaryswitching arrangement 170 may be switched ON, in particular the secondsecondary switching unit 175 and preferably in the second secondarytransistor 176, which allows the input primary current 202 to flowthrough the resonant unit 150 as direct resonant current 251 and throughthe second secondary switching unit 175 as reverse second secondarycurrent 276 as depicted in FIG. 4. This first boost step may comprise afirst rising edge when the second secondary transistor 176 may beswitched ON and a first falling edge when the second secondarytransistor 176 may be switched OFF. At the same time, as it can beobserved on FIG. 33, the direct first primary current 233 decreases andreaches a minimum first primary current, which may be below a firstprimary reference current. Typically, the first primary referencecurrent may be almost zero A.

As the first primary diode 133 turns OFF, the direct first primarycurrent 233 decreases slowly, which may mean with the time rate ofchange of intensity di/dt, of the direct first primary current 233, thereverse recovery energy, E_(RR) for short, may be drastically reduced.

This linearly increase of the reverse second secondary current 276 andthe direct resonant current 251 may be due to the input primary current202, which may be considered as a constant during the first boost stepas shown the equation below:

${I_{FRC}(t)} = \frac{V_{203}*t}{L_{153}}$${I_{DFPC}(t)} \cong {I_{202{\lbrack t_{0}\rbrack}} - \frac{V_{203}*t}{L_{153}}}$V_(DSPV) ≅ V_(203)

As it can be observed on FIG. 33, the second secondary switching unit175, and in particular second secondary transistor 176, turns ON at thezero current and it discharges only its own intrinsic capacitor.

The direct first primary voltage 231 across the resonant unit 150, inparticular across the first capacitor 151, decreases due to a resonancephase. The time rate of change of potential dv/dt across the inductor153, having an inductance L₁₅₃ may be fixed by the natural frequency ofthe resonant unit 150 ω₀, and in particular of the inductor 153 and thefirst and second capacitors 151, 152, having a capacitance C₁₅₁ and C₁₅₂respectively. The natural frequency ω₀ may be given by:

$\omega_{0} = \frac{1}{\sqrt{L_{153}*\left( {C_{151} + C_{152}} \right)}}$with  V_(152)(t) = V_(203) * cos (ω₀t)${{and}\mspace{14mu} {I_{251}(t)}} = {I_{202{\lbrack t_{0}\rbrack}} + {V_{203}*\sqrt{\frac{C_{151} + C_{152}}{L_{252}}}{\sin \left( \omega_{0} \right)}}}$

The direct resonant current 251 rises during the first boost step asmentioned before. As soon as the voltage across the second capacitor152, i.e. the direct second primary voltage 235 measured between theprimary terminal 110 and the common terminal 145, reaches apredetermined ratio of output secondary voltage 203, for example thepredetermined ratio of said output secondary voltage 203 may be maycomprise between 20% and 80%, in particular between 40% and 60% andpreferably between 45% and 55%, the second secondary switching unit 175,in particular the second secondary transistor 176, may be switched off.The duration of [t₁, t₂] may be equal to

$\frac{\pi}{3*\omega_{0}}.$

It may be necessary to precise that a particular effect is achieved at apredetermined ratio of said output secondary voltage 203 equal to 50%more or less 1 percentage point.

When the predetermined ratio of said output secondary voltage 203 may bereached, the second secondary transistor 176 may be switched OFF, thedirect resonant current 251, flowing from the primary terminal 110 tothe common terminal 145 through the resonant unit 150, reaches a maximumdirect resonant current 251 maximum and may be equal at:

$I_{153_{\lbrack t_{2}\rbrack}} = {I_{L_{111}{\lbrack t_{0}\rbrack}} + {\sqrt{\frac{C_{151} + C_{152}}{L_{153}}}*V_{203}*\frac{\sqrt{3}}{2}}}$

Shortly after, a direct first primary voltage 231 measured between theprimary terminal 110 and the secondary terminal 190, in particularacross the first capacitor 151 of the resonant unit 150, and a directsecond primary voltage 235 measured between the primary terminal 110 andthe common terminal 145, in particular across the second capacitor 152of the resonant unit 150, decrease as illustrated in FIG. 5. Thesedirect first primary voltage 231 and direct second primary voltage 235decrease or in others words theses first and second capacitors 151, 152discharge increase linearly a reverse second secondary current 276flowing through the second secondary transistor 176 and the directresonant current 251 flowing through the resonant unit 150 and inparticular in the inductor 153 until reaching a maximum direct resonantcurrent 251 as on FIGS. 6 and 33.

At first falling edge, the second secondary switching unit 175, inparticular the second secondary transistor 176, turns OFF and its timerate of change of potential may be controlled by its intrinsiccharacteristic, and in particular its gate resistor. In others words,the time rate of change of potential of the second secondary transistor176 may be adjusted by the gate resistors to reach a predeterminedvalue.

The choice to switch OFF the second secondary switching unit 175, inparticular the second secondary transistor 176, at a predetermined ratioof said output secondary voltage 203 on the first falling edge, reducesthe resonant current in the resonant unit 150 and the primary andsecondary switching arrangements 130, 170. Thus the power losses insidethe second secondary switching unit 175, in particular inside the secondsecondary transistor 176, may be optimized, to minimize their thermalcycling.

Unlike ordinary transistors working in hard switching, the secondsecondary switching unit 175 of the power converter 100 may not apply afast time rate of change across the load due to the inductor 153. Assoon as, a direct second secondary voltage 275 across the secondsecondary switching unit 175 reaches the output secondary voltage 203,the first secondary switching unit 171, in particular the firstsecondary diode 173, turns ON, and the direct resonant current 251flowing across the inductor begins to decrease as on FIG. 6.

The first falling edge may last around 40 ns in this case and the directresonant current 251 reaches the maximum direct resonant current 251 aspreviously mentioned.

The resonance phase, already mentioned, in on going with the initialcondition I₁₅₃ _([t2]) and V₂₃₅=50%*V₂₀₃ where:

${V_{251}(t)} = {V_{203}\left( {1 - {\cos \left( {{\omega_{0}*t} - \frac{\pi}{3}} \right)}} \right)}$and   ${I_{251}(t)} = {I_{202{\lbrack t_{0}\rbrack}} - {V_{203}*\sqrt{\frac{C_{151} + C_{152}}{L_{153}}}*\left( {\sin \left( {{\omega_{0}*t} - \frac{\pi}{3}} \right)} \right)}}$

To resume the first boost step, the direct first primary current 233flows through the first primary switching unit 131 from the primaryterminal 110 to the secondary terminal 190 and reaches a minimum firstprimary current, the direct second primary voltage 235 measured betweenthe primary terminal 110 and the common terminal 145 drops, the directresonant current 251 flows from the primary terminal 110 to the commonterminal 145 through the resonant unit 150 and reaches a maximum directresonant current 251, and a reverse second primary current 236 flowsthrough the second primary switching unit 135 from the primary terminal110 to the common terminal 145 and may be different than a referencereverse second primary current 236.

FIG. 7 represents the end of the first boost step and the beginning of afirst boost intermediate, where the first and second capacitors 151, 152may be loaded via the first secondary switching unit 171, in particularvia the first secondary diode 173. At this moment, the direct resonantcurrent 251 may be equal to the input primary current 202 of the primaryinductor 111 at the first rising edge and the direct second primaryvoltage 235 across the second capacitor 152 may be equal to zero. Thesecond primary switching unit 135, in particular the second primarydiode 137 in the second primary transistor 136 turns ON, as depicted inFIG. 8, with the minimum direct second primary current 237. The durationof this phase [t₃, t₄] may be equal to

$\frac{\pi}{3*\omega_{0}},$

as it can be observed on FIG. 33.

This period between the first boost step and a second rising edge of asecond boost step may be called first boost intermediate step. In thisfirst boost intermediate step the direct first primary current 233reaches a reference first primary current, the direct second resonantvoltage reaches a reference direct second resonant voltage, the directresonant current 251 drops and the reverse second primary current 236reaches a minimum reverse second primary current 236 as shown on FIG.33.

On FIG. 9, the second primary switching unit 135 may be turned ON andthe second primary diode 137 and the second primary transistor 136 mayconduct at the same time under zero voltage since the threshold voltageof the second primary diode 137 compensates the threshold voltage of thesecond primary transistor 136. Thus, the turn-on energy loss of thesecond primary switching unit 135 may be almost nil or negligible.

Always on FIG. 9, the direct resonant current 251 begins to decrease, asshown on FIG. 33, due to the conduction of the first secondary switchingunit 171, in particular the first secondary diode 173, and the secondprimary switching unit 135, in particular the second primary diode 137,and the energy of resonant unit 150, in particular the energy of theinductor 153, may be transferred towards the secondary terminal 190.

Rule below gives the time rate of change of intensity (di/dt) flowsthrough the first secondary diode 173:

${I_{MDH}(t)} = {{I_{251}(t)} \cong {I_{202{\lbrack t_{0}\rbrack}} - \frac{V_{203}*t}{L_{153}}}}$

As soon as the second primary diode 137 conducts, the second primarytransistor 136 may be switched ON and a second boost step begins and maycomprise a second rising edge when the second primary transistor 136 maybe switched ON and a second falling edge when the second primarytransistor 136 may be switched OFF.

As depicted on FIG. 10, the current crossing the second primarytransistor 136, called reverse second primary current 236, grows sincethe direct resonant current 251 decreases to reach the input primarycurrent value at the first rising edge as shown the rule below:

${I_{236}(t)} = {{I_{202{\lbrack t_{0}\rbrack}} - {I_{251}(t)}} \cong {+ \frac{V_{203}*t}{L_{153}}}}$

As soon as the reverse second primary current 236 reaches the firstprimary current value at the first rising edge, the second boost stepruns and the second primary transistor 136 time conduction may be givenby the control command corresponding to the duty cycle D, around 50%, asillustrated on FIG. 11.

On FIG. 12, the end of the duty cycle D may be represented and thesecond primary switching unit 135, in particular the second primarytransistor 136, may be turned OFF, and the turn-off energy, E_(OFF) forshort, loss may be reduced because the current flows through the firstand second capacitors 151, 152 as shown on the same figure.

At the end of the second boost step, i.e. at the second falling edge,the second primary transistor 136 turns OFF, the direct second primarycurrent 237 dropping may be a combination between the tail current froma bipolar transistor of the second primary transistor 136 and the firstand second capacitors 152:

$\frac{dv}{{dt}_{OFF}} = {\frac{I_{202{\lbrack t_{0}\rbrack}} - I_{{TAIL}_{IGBT}}}{C_{151} + C_{152}}*10^{- ó}\left( {{V/\mu}s} \right)}$

In fact the time rate of change of potential

$\frac{dv}{{dt}_{OFF}}$

may be dependent of the primary switching unit technology, in particularof the first primary switching unit 131 and second primary switchingunit 135 and their junction temperatures. When the junction temperaturegrowths, the tail current increases. This loss will be described later.

FIG. 33 resumes the second boost step wherein the second primaryswitching unit 135 may be switched ON and the direct first primarycurrent 233 stays at said reference first primary current, the directsecond resonant voltage stays at said reference direct second resonantvoltage, the direct resonant current 251 reaches a reference directresonant current 251 and the reverse second primary current 236 reachesa maximum reverse second primary current 236. The second boostintermediate step may be also resumed and described how the direct firstprimary current 233 stays at said reference first primary current, thedirect second resonant voltage increases, direct resonant current 251stays at said reference direct resonant current 251 and the reversesecond primary current 236 drops.

As soon as, the first and second capacitors 151, 152 may be charged bythe primary current, FIG. 13, a third boost step starts and the firstsecondary switching unit 171, in particular the first secondarytransistor 172, turns ON: this may be the third rising edge. In thepresent power converter 100, the conduction of the first secondaryswitching unit 171, in particular of the first secondary transistor 172,discharges the first and second capacitors 151, 152 and the currentflows through the inductor as shown the 13. [t8,t9] duration shows thissequence. Thus the primary switching unit turns ON always at zerovoltage.

Shortly after the charge of the first and second capacitor 152, thefirst primary switching unit 131, in particular the first primary diode133, conducts as represented on FIG. 13.

On a third falling edge, the first secondary switching unit 171, inparticular the first secondary transistor 172, may be switched OFF andthe inductor may be discharged through the first primary switching unit131, in particular through the first primary diode 133, and the secondsecondary switching unit 175, in particular through the second secondarydiode 177 as depicted on FIG. 16.

In this third boost step, the first secondary switching unit 171 may beswitched ON and the direct first primary current 233 reaches a maximumfirst primary current, the direct second resonant voltage reaches amaximum direct second resonant voltage, the direct resonant current 251flows from the primary terminal 110 to the secondary terminal 190through the resonant unit 150 and the reverse second primary current 236reaches said reference reverse second primary current 236.

Between the third falling edge and a forth rising edge, a third boostintermediate step occurs wherein the direct first primary current 233stays at said maximum first primary current, the direct second resonantvoltage stays at said maximum direct second resonant voltage, the directresonant current 251 flows from the primary terminal 110 to thesecondary terminal 190 through the resonant unit 150 and the reversesecond primary current 236 stays at said reference reverse secondprimary current 236 as shown in FIG. 33.

FIG. 33 shows also a fourth boost step wherein said first primaryswitching unit 131 may be switched ON and the direct first primarycurrent 233 stays at said maximum first primary current, the directsecond resonant voltage stays at said maximum direct second resonantvoltage, the direct resonant current 251 flows from the primary terminal110 to the secondary terminal 190 through the resonant unit 150 and thereverse second primary current 236 stays at said reference reversesecond primary current 236.

The first primary switching unit 131, in particular the first primarydiode 133, conduct during a fourth boost intermediate step wherein thedirect first primary current 233 stays at said maximum first primarycurrent, the direct second resonant voltage stays at said maximum directsecond resonant voltage, the direct resonant current 251 flows from theprimary terminal 110 to the secondary terminal 190 through the resonantunit 150 and the reverse second primary current 236 stays at saidreference reverse second primary current 236, as shown on FIG. 17 andthe period phase may be completed.

Also, during the first boost step and the third boost step, the powerlosses may be due to the conduction time of the first secondaryswitching unit 171 and second secondary switching unit 175, discharge ofits intrinsic capacitors and the turn off energy, as shown the equationbelow:

$\begin{matrix}{{PM_{F_{S},P_{0},n}} = {F_{S}*R_{dson}*n}} \\{{*\left\lbrack {\int_{0}^{\frac{P_{0}*L_{153}}{V_{203}*V_{201}}}{\left( \frac{V_{203}t}{L_{153}n} \right)^{2}{dt}}} \right.}} \\{{{+ {\overset{\frac{}{3\omega_{0}}}{\int\limits_{0}}{\left( {\frac{P_{0}}{V_{201}*n} + {\frac{V_{203}}{n}\sqrt{\frac{C_{151} + C_{152}}{L_{153}}}{\sin \left( {\omega_{0}t} \right)}}} \right)^{2}dt}}} + F_{S}}} \\{{{*{\int_{0}^{V_{A}}{Coss_{V_{203}}*V_{203}dV_{203}}}} + F_{S}}} \\\left. {{*{E_{OFF}\left( {\frac{P_{0}}{V_{201}n} + {\frac{V_{203}}{n}\sqrt{\frac{C_{151} + C_{152}}{L_{153}}}*\frac{\sqrt{3}}{2}}} \right)}},R_{G}} \right\rbrack\end{matrix}$

According to the control sequences shown by FIG. 33 and according to theresonant circuit 150 by FIG. 2, the dynamical power losses .i.e at theturn ON E_(ON1) of the primary switching unit 130 and the turn OFFE_(RR1) of the diode function, are nil due to the soft switching and thedynamical power losses of the turn OFF E_(OFF1) of the primary switchingunit 130 are reduced by a ratio close to two, due to the soft switching.Thus, the power losses in conduction and the reduced dynamical powerlosses, E_(ON1)=E_(RR1)=0 and E_(OFF1)/2 mathematically expressed,involve the possibility to let working the semiconductor switch 130 at aswitching frequency well beyond the usual switching frequency which isusually close to 2 or 3 kHz for 1.7 KV IGBTs.

For applications working at 110 A and having a switching frequency of 13kHz, which means 4.5 times higher than the usual frequency, the joulespower losses in the primary switching unit 130, IGBT for example, areequal to 165 W for the static conduction power losses and 377 W for thedynamical switching power losses. Actually, the turn OFF E_(OFF1) of theprimary switching unit 130 in this present invention is divided by twocompared to E_(OFF1) from the hard switching value.

The dynamical power losses of the turn ON E_(ON2) and the turn OFFE_(RR2) of the diode function of the secondary switching unit 170 arenil because the resonant circuit 150 removes these dynamical powerlosses leading at the soft switching. The conduction power losses arelow because the secondary switching unit 170 is flowing through by acurrent for a time ratio of the switching period close to 1/50, as shownin FIG. 33 that highlights the current flowing ILR through the inductor153 that are the same current flowing through the secondary switchingunit 170, and in particular are the same current flowing IML through thesecond secondary transistor 176 during the sequence [t2;t0]. Typically,the conduction losses are close to 6 W for a current of 110 A and thedynamical losses at the turn OFF E_(OFF2) of the secondary switchingunit 170 during the hard switching are equal to 51 W.

Therefore, the power joules losses in the primary switching unit 130 areequal to 542 W, either 9.5 times higher than the secondary switchingunit 170, which is 57 W.

For comparison, the primary switching unit 130, without the secondaryswitching unit 170 associated with its resonance circuit 150, wouldreach a power losses close to 2323 W at 13 kHz which is giving by 165 Wof conduction power losses and 2158 W for the dynamical power lossesduring the commutations, i.e. (58 mJ+53 mJ+55 mJ)×13 kHz. Such a devicecannot work because the skilled people in the art know that the junctiontemperature would be well beyond the limits allowed by IGBTssemiconductor components.

Thus, the present invention, which is characterized by primary dynamicalswitching losses of the primary switching unit 130 greater than thesecondary dynamical switching losses of the secondary switching unit170, provides the possibility to reduce, on the one hand, the totalpower losses of the power converter by a ratio equal to four compared tothe state of the art and on the other hand, it allows to increase theswitching frequency by a ratio equal at four, for the IGBT 1.7 kV.

The realization of the primary switching unit 130 from the materialconstituting the semiconductor of the secondary switching unit 170 hasno economic sense because price of the function of the primary switchingunit 130, would be estimated to 810 €, i.e. (450 mm²/36 mm²) 12.5secondary switching unit 170 at 1.8 €, which is compared to the presentinvention estimated about 92 €, i.e. 450 mm² of the primary switchingunit 130 at 0.06 € and 36 mm² of the secondary switching unit 170 at 1.8€ adds up to 92 €; that is a ratio of eight.

The previously analysis and comparison can be made with the 1.2 kV IGBT,or other kind of components comprised in the primary switching unit 130,having lower conduction power losses than a secondary switching unit 170and having higher power dynamical power losses than the secondaryswitching unit 170.

FIG. 34 shows the maximum delta temperature ΔT=P_(MOSFET)×R_(th(j-h))versus the number n of SiC power MOSFET connected in parallel. To avoidaging of first secondary transistor 172 and/or the second secondarytransistor 176 due to the thermal cycling, a maximum differentialtemperature equal to 20° C. may be considered. In this case, TO247package may be enough to make a power converter 100 with an output powerequal to 63 kW.

Although the direct first secondary voltage 271 and direct secondsecondary voltage 275 may be almost equal to 950V, the first secondarytransistor 172 and second secondary transistor 176 may be used due totheir high immunity against the cosmic rays.

The zero voltage switching method used in the power converter 100,removes the EON energy. However, E_(OFF1) and E_(RR) contribute to thepower losses. E_(RR) of the second primary diode 137 in parallel of thesecond primary transistors 136, keeps its recovery charges when thesecond primary transistor 136 may be ON between the second rising edgeand the second falling edge. It may be at the second falling edge, whenthe direct second primary voltage 235 rises, that its charges will beevacuated, involving the power losses in the second primary diode 137.

However, the E_(RR) energy may be low versus the E_(OFF). E_(OFF)corresponds to the recombination of the charge of the bipolar transistorof the first primary transistor 132 and/or the second primary transistor136. This phenomenon may be known as tail current. FIG. 34 shows theenergy sum of E_(OFF) and E_(RR) E_(total) when the first secondarytransistor 172 or second primary transistor 136 turns OFF.

As the current may be measured by an external probe of the powerconverter, tail current may be the summation of the direct secondprimary current 237 and reverse second primary current 236.

Power converter 100 may be used on all the applications needing highintegration, high reliability and high efficiency, with many thermalcycles, in an aggressive market.

Description of a Buck Process

The description hereafter shows the sequences when the power converter100 works like a Buck power converter 100 and implementing a buckprocess, which may mean that an energy source may be connected to thesecondary terminal 190 and the common terminal 145 and a load may beconnected to the primary terminal 110 and the common terminal 145. Forreading convenience and better understanding, the switching and resonantunit 150 s, which may be OFF, may be dotted and others, which may be ON,may be draw with solid line.

In the present invention, by “OFF” and “no current” we mean that almostno current flows, a current lower or equal than 1 mA or the flowingcurrent may be not intended. On other side, by “ON” we mean that acurrent flows, a current higher than 1 mA or the flowing current may beintended.

Also, the terms direct and reverse may be just used for readingconvenience and may indicate upward and downward respectively or rightand left respectively in the direction of the figure.

Further, by the terms “load”, we mean all electronic/electrical deviceswhich transform or consume electrical energy like a motor, a battery, asupercapacitor or a resistor, but not limited thereto, and by “source”we mean all energy suppliers which supply electrical energy like amotor, a battery, or a supercapacitor for instance, but not limitedthereto.

During an initial step, a direct second primary current 237 flowsthrough the primary inductor 111 and the primary switching arrangement130 from the energy source to the primary terminal 110. In particular,the second secondary current flows to the secondary terminal 190 throughthe primary inductor 111 and at least one of a primary switching unitmay comprise in the primary switching arrangement 130 and preferably inthe second primary switching unit 135. No current flows in the othersunits, which may mean no current flows through a first primary switchingunit 131 may comprise in the primary switching unit, the resonant unit150 and the secondary switching unit.

As depicted in FIG. 21), a direct second primary current 237 flowsthrough the second primary diode 137 of the first primary switching unit131 from the secondary terminal 190 to the primary terminal 110.

After this initial step, a first buck step occurs wherein the secondaryswitching unit may be switched ON, in particular the first secondaryswitching unit 171 and preferably in the first secondary transistor 172,which allows the second secondary current to flow through the resonantunit 150 as reverse direct resonant current 251 and through the firstsecondary switching unit 171 as reverse first secondary current 272 asdepicted in FIG. 22). This first buck step may comprise a first risingedge when the first secondary transistor 172 may be switched ON and afirst falling edge when the first secondary transistor 172 may beswitched OFF. At the same time, the direct second primary current 237decreases and reaches a minimum second primary current which may bebelow a second primary reference current. Typically, the second primaryreference current may be almost 0 A.

As the second primary diode 137 turns OFF, the direct second primarycurrent 237 decreases slowly, which may mean with the time rate ofchange of current di/dt, of the direct second primary current 237, thereverse recovery energy E_(RR) may be drastically reduced.

This linearly increase of the reverse first secondary current 272 andthe reverse direct resonant current 251 may be due to the secondarycurrent, which may be considered as a constant during the first buckstep. The equation, shown previously, apply also in the buck process,but the difference may be that the V₂₀₃ replace V₂₀₁ and V₂₉₃ replaceV₂₉₁.

The first secondary switching unit 171, and in particular firstsecondary transistor 172, turns ON at the zero current and it dischargesonly its own intrinsic capacitor.

The reverse first primary voltage 102 across the resonant unit 150, inparticular across the first and second capacitor 152 s, decreases due toa resonance phase. The time rate of change of potential dv/dt across theinductor may be fixed by the natural frequency of the resonant unit 150ω₀, and in particular of the inductor and the first and second capacitor152 s.

The reverse direct resonant current 251 rises during the first buck stepas mentioned before. As soon as the voltage across the first capacitor151, i.e. the direct first resonant voltage measured between the primaryterminal 110 and the common terminal 145, reaches a predetermined ratioof second primary voltage, for example the predetermined ratio of saidsecond primary voltage may be may comprise between 20% and 80%, inparticular between 40% and 60% and preferably between 45% and 55%, thefirst secondary switching unit 171, in particular the first secondarytransistor 172, may be switched off. This duration may be also equal toπ/3*ω₀. It may be necessary to precise that a particular effect may beachieved at a predetermined ratio of said second primary voltage equalto 50% more or less 1 percentage point.

When the predetermined ratio of said second primary voltage may bereached, the first secondary transistor 172 may be switched OFF, thereverse direct resonant current 251, flowing from the secondary terminal190 to the common terminal 145 through the resonant unit 150, reaches adirect resonant current 251 maximum.

Shorty after, a direct first primary voltage 231 measured between theprimary terminal 110 and the secondary terminal 190, in particularacross the first capacitor 151 of the resonant unit 150, and a directsecond primary voltage 235 measured between the primary terminal 110 andthe common terminal 145, in particular across the second capacitor 152of the resonant unit 150, decrease as illustrated in FIG. 23). Thesedirect first primary voltage 231 and direct second primary voltage 235decrease or in others words theses first and second capacitor 152 sdischarge increase linearly a reverse first secondary current 272flowing through the first secondary transistor 172 and the reversedirect resonant current 251 flowing through the resonant unit 150 and inparticular in the inductor until reaching a maximum direct resonantcurrent 251 as on FIG. 24)

At first falling edge, the first secondary switching unit 171, inparticular the first secondary transistor 172, turns OFF and its timerate of change of potential may be controlled by its intrinsiccharacteristic, and in particular its gate resistor. In others words,the time rate of change of potential of the first secondary transistor172 may be adjusted by the gate resistors to reach a predeterminedvalue.

The choice to switch OFF the first secondary switching unit 171, inparticular the first secondary transistor 172, at a predetermined ratioof said output secondary voltage on the first falling edge, reduces thereverse resonant current in the resonant unit 150 and the primary andsecondary switching unit. Thus the power losses inside the firstsecondary switching unit 171, in particular inside the first secondarytransistor 172, may be optimized, to minimize their thermal cycling.

Unlike ordinary transistors working in hard switching, the firstsecondary switching unit 171 of the power converter 100 may not apply afast time rate of change across the load due to the inductor. As soonas, a direct first secondary voltage 271 across the first secondaryswitching unit 171 reaches the second primary voltage, the secondsecondary switching unit 175, in particular the second secondary diode177, turns ON, and the reverse direct resonant current 251 flowingacross the inductor begins to decrease as on FIG. 4).

The first falling edge may last around 40 ns in this case and the secondresonant current reaches the maximum second resonant current aspreviously mentioned.

FIG. 25) represents the end of the first buck step and the beginning ofa first buck intermediate, where the first and second capacitor 152 smay be loaded via the second secondary switching unit 175, in particularvia the second secondary diode 177. At this moment, the second resonantcurrent may be equal to the reverse primary current of the primaryinductor 111 at the first rising edge and the direct first primaryvoltage 231 across the first capacitor 151 may be equal to zero. Thefirst primary switching unit 131, in particular the first primary diode133 in the first primary transistor 132 turns ON, as depicted in FIG.26), with the minimum direct first primary current 233. The duration ofthis phase may be also equal to

$\frac{\pi}{3*\omega_{0}}.$

On FIG. 27), the first primary switching unit 131 may be turned ON andthe first primary diode 133 and the first primary transistor 132 mayconduct at the same time under zero voltage since the threshold voltageof the first primary diode 133 compensates the threshold voltage of thefirst primary transistor 132. Thus, the turn on energy loss of the firstprimary switching unit 131 may be almost nil or negligible.

Always on FIG. 27), the first reverse resonant current begins todecrease due to the conduction of the second secondary switching unit175, in particular the second secondary diode 177, and the first primaryswitching unit 131, in particular the first primary diode 133, and theenergy of resonant unit 150, in particular the energy of the inductor,may be transferred towards the primary terminal 110.

Rule below gives the time rate of change of current di/dt flows throughthe second secondary diode 177.

As soon as the first primary diode 133 conducts, the first primarytransistor 132 may be switched ON and a second boost step begins and maycomprise a second rising edge when the first primary transistor 132 maybe switched ON and a second falling edge when the first primarytransistor 132 may be switched OFF.

As depicted on FIG. 28), the current crossing the first primarytransistor 132, called reverse first primary current, grows since thereverse direct resonant current 251 decreases to reach the secondsecondary current value at the first rising edge as shown the rulebelow.

As soon as the reverse first primary current reaches the secondsecondary current value at the first rising edge, the second buck stepruns and the first primary transistor 132 time conduction may be givenby the control command corresponding to the duty cycle D as illustratedon FIG. 29).

On FIG. 30), the end of the duty cycle D may be represented and thefirst primary switching unit 131, in particular the first primarytransistor 132, may be turned OFF, and the turn off energy loss may bereduced because the current flows through the first and second capacitor152 s as shown on the same figure.

At the end of the second buck step, i.e. at the second falling edge, thefirst primary transistor 132 turns OFF, the direct first primary current233 dropping may be a combination between the tail current from abipolar transistor of the first primary transistor 132 and the first andsecond capacitor 152 s.

In fact the time rate of change of potential

$\frac{dv}{{dt}_{OFF}}$

may be dependent of the primary switching unit technology, in particularof the first primary switching unit 131 and second primary switchingunit 135 and their junction temperatures. When the junction temperaturegrowths, the tail current increases. This loss will be described later.

As soon as, the first and second capacitor 152 s may be charged by thesecondary current, FIG. 31), a third buck step starts and the secondsecondary switching unit 175, in particular the second secondarytransistor 176, turns ON: this may be the third rising edge. In thepresent power converter 100, the conduction of the second secondaryswitching unit 175, in particular of the second secondary transistor176, discharges the first and second capacitors 152 and the currentflows through the inductor as shown the 31). Thus the primary switchingunit turns ON always at zero voltage.

Shortly after the charge of the first and second capacitor 152, thesecond primary switching unit 135, in particular the second primarydiode 137, conducts as represented on FIG. 32).

On a third falling edge, the second secondary switching unit 175, inparticular the second secondary transistor 176, may be switched OFF andthe inductor may be discharged through the second primary switching unit135, in particular through the second primary diode 137, and the firstsecondary switching unit 171, in particular through the first secondarydiode 173 as depicted on FIG. 34).

The second primary switching unit 135, in particular the second primarydiode 137, conduct as shown on FIG. 35) and the period phase may becompleted.

Also, during the first buck step and the third buck step, the powerlosses may be due to the conduction time of the second secondaryswitching unit 175 and first secondary switching unit 171, discharge ofits intrinsic capacitors and the turn off energy, as shown the equationabove.

1. A boost or buck power converter for converting an input primary powerhaving an input primary voltage and an input primary current into aoutput secondary power having an output secondary voltage and outputsecondary current or an input secondary power having said inputsecondary voltage and an input secondary current into an output primarypower having an output primary voltage and second primary currentrespectively; said power converter including at least one: primaryswitching arrangement comprising at least one primary switching unithaving a primary dynamical switching loss; secondary switchingarrangement comprising at least one secondary switching unit having asecondary dynamical switching loss; and, resonant unit connecting saidat least one primary switching unit and said at least one secondaryswitching unit; said primary dynamical switching loss being higher thansaid secondary dynamical switching loss.
 2. The power converteraccording to claim 1, wherein said resonant unit may comprise at leastone inductor and one capacitor.
 3. The power converter according toclaim 1, wherein said primary switching arrangement may comprise atleast a first primary switching unit and a second primary switching unitand/or wherein said secondary switching arrangement may comprise atleast a first secondary switching unit and a second secondary switchingunit.
 4. The power converter according to claim 1, wherein a dimensionof primary switching arrangement is larger than a dimension of thesecondary switching arrangement.
 5. The power converter according toclaim 1, wherein said at least one primary switching unit is a GateTurn-OFF thyristor, Insulated-Gate Bipolar Transistor, Field EffectTransistor and/or Metal-Oxide-Semiconductor Field-Effect Transistorand/or said at least one secondary switching unit is a Field EffectTransistor and/or a Metal-Oxide-Semiconductor Field-Effect Transistor.6. The power converter according to claim 1, comprising at least one:primary terminal: said primary terminal is configured to connect aprimary electrical component 901 to said power converter; secondaryterminal: said secondary terminal is configured to connect a secondaryelectrical component 902 to said power converter; common terminal saidcommon terminal is connected to said primary electrical component 901 tosaid secondary electrical component 902; said primary switchingarrangement is connected to said primary terminal, to said secondaryterminal and to said common terminal, and said secondary switchingarrangement is to connected to said secondary terminal and to saidcommon terminal.
 7. A boost or buck process for converting a inputprimary power having a input primary voltage and a input primary currentinto a output secondary power having a output secondary voltage andoutput secondary current or a input secondary power having said inputsecondary voltage and a input secondary current into a output primarypower having a output primary voltage and second primary currentrespectively; said boost or buck process comprising at least a step of:providing a power converter according to claim 1; switching OFF of atleast one secondary switching unit when a voltage across said at leastone primary switching unit reaches a predetermined ratio of said outputsecondary voltage or said output primary voltage respectively.
 8. Theboost or buck process according to claim 7, wherein said predeterminedratio of said output secondary voltage or said output primary voltagerespectively is may comprise between 20% and 80%, in particular between40% and 60% and preferably between 45% and 55%.
 9. The boost or buckprocess according to claim 7, wherein said voltage across said at leastone primary switching unit (131, 135) is a voltage across said at leastone capacitor.
 10. The boost or buck process according to claim 7,wherein said at least one secondary switching unit comprises anintrinsic characteristic controlling the time of said switching OFFstep.
 11. The boost or buck process according to claim 10, wherein saidintrinsic characteristic is a gate resistor of said at least onesecondary switching unit.
 12. The buck process according to claim 7,wherein said voltage across said at least one primary switching unitreaches a voltage between said secondary terminal and said primaryterminal.
 13. The boost process according to claim 7, wherein saidvoltage across said at least one primary switching unit reaches avoltage between said primary terminal and said common terminal.
 14. Theboost or buck process according to claim 7, wherein a current of saidresonant unit reaches a maximum resonant current.
 15. The boost or buckprocess according to claim 7, comprising a switch ON step of said atleast one primary switching unit when said voltage across said at leastone primary switching unit reaches a minimum voltage.
 16. The powerconverter according to claim 2, wherein said primary switchingarrangement may comprise at least a first primary switching unit and asecond primary switching unit and/or wherein said secondary switchingarrangement may comprise at least a first secondary switching unit and asecond secondary switching unit.
 17. The power converter according toclaim 16, wherein a dimension of primary switching arrangement is largerthan a dimension of the secondary switching arrangement.
 18. The powerconverter according to claim 17, wherein said at least one primaryswitching unit is a Gate Turn-OFF thyristor, Insulated-Gate BipolarTransistor, Field Effect Transistor and/or Metal-Oxide-SemiconductorField-Effect Transistor and/or said at least one secondary switchingunit is a Field Effect Transistor and/or a Metal-Oxide-SemiconductorField-Effect Transistor.
 19. The power converter according to claim 18,comprising at least one: primary terminal: said primary terminal isconfigured to connect a primary electrical component 901 to said powerconverter; secondary terminal: said secondary terminal is configured toconnect a secondary electrical component 902 to said power converter;common terminal: said common terminal is connected to said primaryelectrical component 901 to said secondary electrical component 902;said primary switching arrangement is connected to said primaryterminal, to said secondary terminal, and to said common terminal, andsaid secondary switching arrangement is to connected to said secondaryterminal and to said common terminal.
 20. The boost or buck processaccording to claim 8, wherein said voltage across said at least oneprimary switching unit (131, 135) is a voltage across said at least onecapacitor.